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Antenna Simulation in LTspice
I am working on a simulation for a loop antenna in LTspice and I can't
figure out why the signal strength features are what they are. The model uses a pair of loosely coupled inductors to model the transmitter and antenna loop with a separate pair of tightly coupled inductors to model the coupling transformer. A cap on the primary circuit is the tuning cap and a cap on the secondary is parasitic effects of the circuit board leading to the inputs on the IC. There is a resonance near the frequency I would expect, but it is not so close actually. I can't figure why it is about 5% off. There is a second resonance fairly high up that I can't figure at all. None of the component values seem to combine appropriately to produce this peak. When looking at the tuning capacitor voltage there is an anti-resonance that is exactly at the frequency corresponding to the secondary resonance with the transformer and the parasitic capacitance. That makes sense to me, but it is pretty much the only part that jibes with what I can figure out. I have uploaded a zip file with the schematic and a measurement file. http://arius.com/temp/Antenna_trans_LTspice.zip -- Rick |
Antenna Simulation in LTspice
I expect if you reflect the CT secondary stuff (don't forget Lsec) back to
the primary, your answer will appear. Offhand I can't reason out which sum of L and C makes the resonance, but it's a four pole series-parallel resonant circuit, analysis should lay it bare. Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com "rickman" wrote in message ... I am working on a simulation for a loop antenna in LTspice and I can't figure out why the signal strength features are what they are. The model uses a pair of loosely coupled inductors to model the transmitter and antenna loop with a separate pair of tightly coupled inductors to model the coupling transformer. A cap on the primary circuit is the tuning cap and a cap on the secondary is parasitic effects of the circuit board leading to the inputs on the IC. There is a resonance near the frequency I would expect, but it is not so close actually. I can't figure why it is about 5% off. There is a second resonance fairly high up that I can't figure at all. None of the component values seem to combine appropriately to produce this peak. When looking at the tuning capacitor voltage there is an anti-resonance that is exactly at the frequency corresponding to the secondary resonance with the transformer and the parasitic capacitance. That makes sense to me, but it is pretty much the only part that jibes with what I can figure out. I have uploaded a zip file with the schematic and a measurement file. http://arius.com/temp/Antenna_trans_LTspice.zip -- Rick |
Antenna Simulation in LTspice
On 2/27/2013 11:40 PM, Tim Williams wrote:
I expect if you reflect the CT secondary stuff (don't forget Lsec) back to the primary, your answer will appear. Offhand I can't reason out which sum of L and C makes the resonance, but it's a four pole series-parallel resonant circuit, analysis should lay it bare. Are you sure about this? When you say to reflect the secondary back to the primary that means the primary inductance would be doubled? I believe the coupling of the two coils means they are one and the same for the purposes of the circuit analysis, no? You can't reason which sum of L and C makes the resonance and I can't either. The calculation is off by about 5% and I can't explain that. I can explain a null at about 290 kHz. That is the resonance of the secondary with the secondary capacitance. I can't explain the other peak at 363 kHz at all. A higher frequency would imply a smaller L and/or C. How do you combine them to produce that? Consider the two caps to be in series??? -- Rick |
Antenna Simulation in LTspice
"rickman" wrote in message
... A higher frequency would imply a smaller L and/or C. How do you combine them to produce that? Consider the two caps to be in series??? Sure. If you bring the 10p over to the primary, it looks like 10p * (30m / 5u), or whatever the ratio was (I don't have it in front of me now), in parallel with the primary. (I misspoke earlier, you can safely ignore Ls, because k = 1. There's no flux which is not common to both windings.) Inductors effectively in parallel also increase the expected resonant frequency. If you have this, .. L1 .. +-----UUU--+------+------+ .. | + | | | .. ( Vsrc ) === C R 3 L2 .. | - | 3 .. | | | | .. +----------+------+------+ .. _|_ GND You might expect the resonant frequency is L2 + C, but it's actually (L1 || L2) = Leq. If L1 is not substantially larger than L2, the resonant frequency will be pulled higher. Incidentally, don't forget to include loss components. I didn't see any explict R on the schematic. I didn't check if you set the LTSpice default parasitic ESR (cap), or DCR or EPR (coil) on the components. Besides parasitic losses, your signal is going *somewhere*, and that "where" consumes power! The actual transmitter is most certainly not a perfect current source inductor, nor is the receiver lossless. This simulation has no expression for radiation in any direction that's not directly between the two antennas: if all the power transmitted by the current source is reflected back, even though it's through a 0.1% coupling coefficient, it has to go somewhere. If it's coming back out the antenna, and it's not being burned in the "transformer", it's coming back into the transmitter. This is at odds with reality, where a 100% reflective antenna doesn't magically smoke a distant transmitter, it simply reflects 99.9% back into space. The transmitter hardly knows. In this example, if you set R very large, you'll see ever more voltage on the output, and ever more current draw from Vsrc. You can mitigate this by increasing L1 still further, but the point is, if the source and load (R) aren't matched in some fashion, the power will reflect back to the transmitter and cause problems (in this case, power reflected back in-phase causes excessive current draw; in the CCS case, reflected power in-phase causes minimal voltage generation and little power transmission). Power is always coming and going somewhere, and if you happen to forget this fact, it'll reflect back and zap you in the butt sooner or later! Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com |
Antenna Simulation in LTspice
On Wed, 27 Feb 2013 22:07:51 -0500, rickman wrote:
snip There is a resonance near the frequency I would expect, but it is not so close actually. I can't figure why it is about 5% off. There is a second resonance fairly high up that I can't figure at all. None of the component values seem to combine appropriately to produce this peak. snip Pulling out the old reactance paper, there are a couple of expected interactions using the values present: Around 50KHz (89.42uH+48uH) with 50.42nF (L3+L1) with C1 Around 290KHz (89.42uH+48uH) with 6.25nF (L3+L1) with C2*N^2 Around 360KHz 48uH with 6.25nF L1 with C2*N^2 nL1/nL2=N=25 The mid-resonance is a dip or rejection. What's the issue? RL |
Antenna Simulation in LTspice
On Wed, 27 Feb 2013 22:07:51 -0500, rickman wrote:
I am working on a simulation for a loop antenna in LTspice and I can't figure out why the signal strength features are what they are. The model uses a pair of loosely coupled inductors to model the transmitter and antenna loop with a separate pair of tightly coupled inductors to model the coupling transformer. A cap on the primary circuit is the tuning cap and a cap on the secondary is parasitic effects of the circuit board leading to the inputs on the IC. There is a resonance near the frequency I would expect, but it is not so close actually. I can't figure why it is about 5% off. There is a second resonance fairly high up that I can't figure at all. None of the component values seem to combine appropriately to produce this peak. When looking at the tuning capacitor voltage there is an anti-resonance that is exactly at the frequency corresponding to the secondary resonance with the transformer and the parasitic capacitance. That makes sense to me, but it is pretty much the only part that jibes with what I can figure out. I have uploaded a zip file with the schematic and a measurement file. http://arius.com/temp/Antenna_trans_LTspice.zip Re-read what Tim Williams said. A way to translate what he's saying into your simulation is to include the radiation resistance of the antenna into your simulation, and reduce the coupling -- 1e-6 is probably good enough. I am, frankly, not sure where this is best put in your radiation resistance, but just increasing the series resistance on L3 is probably sufficient; putting it in as a parallel resistance in L3 is probably more accurate, but would be more useful as a way of separating the radiation resistance effect from the winding resistance of L3 (which is, I assume, where your figure comes from). Do you have the ability to measure the Q of your antenna as built, and compare it to the Q calculated from the known L, C, and winding resistance? That should give you a good estimate of the radiation resistance, or at least radiation resistance + other losses. -- My liberal friends think I'm a conservative kook. My conservative friends think I'm a liberal kook. Why am I not happy that they have found common ground? Tim Wescott, Communications, Control, Circuits & Software http://www.wescottdesign.com |
Antenna Simulation in LTspice
On 2/28/2013 6:40 PM, Tim Williams wrote:
wrote in message ... A higher frequency would imply a smaller L and/or C. How do you combine them to produce that? Consider the two caps to be in series??? Sure. If you bring the 10p over to the primary, it looks like 10p * (30m / 5u), or whatever the ratio was (I don't have it in front of me now), in parallel with the primary. (I misspoke earlier, you can safely ignore Ls, because k = 1. There's no flux which is not common to both windings.) Reflecting the capacitance through the transformer changes it by the square of the turns ratio assuming the coupling coefficient is sufficiently high. I am simulating K at 1. This is also true for the inductance, but in the opposite manner. So going from the 25 turn side to the 1 turn side, the effective capacitance is multiplied by 625 and the effective inductance (or resistance) is divided by 625. In fact, in LTspice you indicate the turns ratio by setting the inductance of the two coils by this ratio. I see now that the reflected secondary capacitance is in parallel with the primary, rather than in parallel with the primary capacitor. That explains a lot... I'll have to hit the books to see how to calculate this new arrangement. I found a very similar circuit in the Radiotron Designer's Handbook. In section 4.6(iv)E on page 152 they show a series-parallel combination that only differs in the placement of the resistance in the parallel circuit. It need to be placed inline with the inductor... or is placing it parallel correct since this is the reflected resistance of the secondary? I'll have to cogitate on that a bit. I'm thinking it would be properly placed inline with the capacitor in the reflection since it is essentially inline in the secondary. Either way I expect it will have little impact on the resonant frequency and I can just toss all the resistances simplifying the math. I do see one thing immediately. The null in Vcap I see is explained by the parallel resonance of the secondary cap with the secondary inductor. If you reflect that cap back to the primary in parallel with the primary inductor (resonating at the same frequency) it explains the null in the capacitor C1 voltage I see. C2' (reflected) and L1 make a parallel resonance with a high impedance dropping the primary cap current and voltage to a null. This null is calculated accurately. What I need to do is change the impedance equation from Radiotron to one indicating the voltage at Vout relative to the input signal. I think I can do that by treating the circuit as a voltage divider taking the ratio of the impedance at the input versus the impedance at the primary coil. No? Inductors effectively in parallel also increase the expected resonant frequency. If you have this, . L1 . +-----UUU--+------+------+ . | + | | | . ( Vsrc ) === C R 3 L2 . | - | 3 . | | | | . +----------+------+------+ . _|_ GND You might expect the resonant frequency is L2 + C, but it's actually (L1 || L2) = Leq. If L1 is not substantially larger than L2, the resonant frequency will be pulled higher. I see, L1 and L2 are in parallel because the impedance of Vsrc is very low. That is not the circuit I am simulating however. The loop of the antenna and the loop of the inductor are in series along with the primary capacitor. I'm not sure what the resistor is intended to represent, perhaps transformer losses? The resistance of L1 was added to the simulation model along with the resistance of the secondary coil which you have not shown... I think. It seems to me you have left out the tuning capacitor on the primary. Incidentally, don't forget to include loss components. I didn't see any explict R on the schematic. I didn't check if you set the LTSpice default parasitic ESR (cap), or DCR or EPR (coil) on the components. Besides parasitic losses, your signal is going *somewhere*, and that "where" consumes power! The actual transmitter is most certainly not a perfect current source inductor, nor is the receiver lossless. This simulation has no expression for radiation in any direction that's not directly between the two antennas: if all the power transmitted by the current source is reflected back, even though it's through a 0.1% coupling coefficient, it has to go somewhere. If it's coming back out the antenna, and it's not being burned in the "transformer", it's coming back into the transmitter. This is at odds with reality, where a 100% reflective antenna doesn't magically smoke a distant transmitter, it simply reflects 99.9% back into space. The transmitter hardly knows. Interesting point. My primary goal with this is to simulate the resonance of the tuning so I can understand how to best tune the circuit. In many of the simulations I run the Q ends up being high enough that a very small drift in the parasitic capacitance on the secondary detunes the antenna and drops the signal level. It sounds like there are other losses that will bring the Q much lower. I would also like to have some idea of the signal strength to expect. My understanding is that the radiation resistance of loop antennas is pretty low. So not much energy will be radiated out. No? You make it sound as if in the simulation, even with a small coupling coefficient all the energy from antenna inductor will still couple back into the transmitter inductor regardless of the K value. Do I misunderstand you? It seems to result in the opposite, minimizing this back coupling. Or are you saying that the simulation needs to simulate the radiation resistance to show radiated losses? In this example, if you set R very large, you'll see ever more voltage on the output, and ever more current draw from Vsrc. You can mitigate this by increasing L1 still further, but the point is, if the source and load (R) aren't matched in some fashion, the power will reflect back to the transmitter and cause problems (in this case, power reflected back in-phase causes excessive current draw; in the CCS case, reflected power in-phase causes minimal voltage generation and little power transmission). Power is always coming and going somewhere, and if you happen to forget this fact, it'll reflect back and zap you in the butt sooner or later! Tim Actually, my goal was to build the receiver and I realized that my design would require the largest signal I could get from the antenna. I never realized I would end up having to learn quite so much about antenna design. I've been planning to create a PCB with lots of options so I can test a number of configurations. Nothing about the simulation makes me doubt the utility of this idea. One thing that continues to bug me is that nothing I have seen gives me a hint on how to factor in the distributed capacitance of the antenna shield. I am using RG6 with 16 pF/Ft and likely will end up with 100 foot of coax total. At some point I'll just have to make some measurements and see what the real world does. -- Rick |
Antenna Simulation in LTspice
You'll be much better off simply using the conventional radio approach
than trying to simulate everything, especially when circuit equivalents are nebulous like this. After all, if you can't quite tell what it *should* look like, how would you know if you could implement your model once you've found a satisfactory result? What kind of antenna are you looking at, loop? The first thing to know about a loop is, if it's a very small loop (I'm guessing, at this frequency, it is), its radiation resistance is very low, meaning, you can treat it as a nearly pure inductance (Q 10 I think is typical), and its bandwidth (even with a matched load) will be correspondingly narrow. The nature of the incoming signal could be modeled as a voltage or current source; how doesn't really matter, because it isn't really either, it's a power source that couples in. Again, you don't have voltage without current and vice versa, it's all about power flow, and the matching that allows the power to flow. Since the loop is inductive, your first priority is to resonate it with a capacitor at the desired frequency. This will require a very precise value, and even for a single frequency, may require a variable capacitor to account for manufacturing tolerances. In the AM BCB, a Q of 10 gets you 50-160kHz bandwidth, so you only get a few channels for any given tuning position. And if the Q is higher, you get even fewer. Now that you've got a high Q resonant tank, you can do two things: couple into the voltage across the capacitor, or the current through the inductor. You need only a small fraction of either, because the Q is still going to be large. This can be arranged with a voltage divider (usually the capacitor is split into a huge hunk and a small variable part, e.g., 300pF variable + 10nF, output from across the 10nF), a transformer (a potential transformer across the cap, or a current transformer in series with the inductor), an inductive pickup (the big loop carries lots of volts, but you only need a few, so a much smaller loop can be placed inside the big loop), an impractically large inductor (like in my example circuit, which models radiation resistance as a parallel equivalent), etc. Whatever the case, you need to match transmission line impedance (e.g., 50 ohms) to radiation resistance (whichever series or parallel equivalent you have). Once you get the signal into a transmission line, with a reasonable match (Z ~= Z_line, or alternately, SWR ~= 1), you can do whatever you want with it. Put it into an amplifier (don't forget to match it, too), etc. Yes, you're going to have funny behavior at other frequencies, and if you're concerned about those frequencies, you'll have to choose the coupling circuit and adjustable (or selectable) components accordingly. But for the most part, you completely ignore any frequency that you aren't tuning for, usually enforcing that concept by inserting filters to reject any stragglers. Example: suppose you have a loop of 5uH and need to tune it to 500kHz. It has a reactance of 15.7 ohms. Suppose further it has Q = 20. The ESR (not counting DCR and skin effect) is X_L / Q, or 0.78 ohms; alternately, the EPR is X_L * Q, or 314 ohms. The capacitor required is 20.3nF. If we use a current transformer to match to a 50 ohm line, it needs an impedance ratio of 1:64, or a turns ratio of 1:8. If we use a voltage transformer, it's of course 8:1. (A capacitor divider is unsuitable for resonant impedances less than line impedance, since it can only divide the impedance down. If the inductance were a lot larger, it could be used.) To a rough approximation, a smaller inductive loop, of 1/8 diameter of the larger, I think, would also work. Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com "rickman" wrote in message ... On 2/28/2013 6:40 PM, Tim Williams wrote: wrote in message ... A higher frequency would imply a smaller L and/or C. How do you combine them to produce that? Consider the two caps to be in series??? Sure. If you bring the 10p over to the primary, it looks like 10p * (30m / 5u), or whatever the ratio was (I don't have it in front of me now), in parallel with the primary. (I misspoke earlier, you can safely ignore Ls, because k = 1. There's no flux which is not common to both windings.) Reflecting the capacitance through the transformer changes it by the square of the turns ratio assuming the coupling coefficient is sufficiently high. I am simulating K at 1. This is also true for the inductance, but in the opposite manner. So going from the 25 turn side to the 1 turn side, the effective capacitance is multiplied by 625 and the effective inductance (or resistance) is divided by 625. In fact, in LTspice you indicate the turns ratio by setting the inductance of the two coils by this ratio. I see now that the reflected secondary capacitance is in parallel with the primary, rather than in parallel with the primary capacitor. That explains a lot... I'll have to hit the books to see how to calculate this new arrangement. I found a very similar circuit in the Radiotron Designer's Handbook. In section 4.6(iv)E on page 152 they show a series-parallel combination that only differs in the placement of the resistance in the parallel circuit. It need to be placed inline with the inductor... or is placing it parallel correct since this is the reflected resistance of the secondary? I'll have to cogitate on that a bit. I'm thinking it would be properly placed inline with the capacitor in the reflection since it is essentially inline in the secondary. Either way I expect it will have little impact on the resonant frequency and I can just toss all the resistances simplifying the math. I do see one thing immediately. The null in Vcap I see is explained by the parallel resonance of the secondary cap with the secondary inductor. If you reflect that cap back to the primary in parallel with the primary inductor (resonating at the same frequency) it explains the null in the capacitor C1 voltage I see. C2' (reflected) and L1 make a parallel resonance with a high impedance dropping the primary cap current and voltage to a null. This null is calculated accurately. What I need to do is change the impedance equation from Radiotron to one indicating the voltage at Vout relative to the input signal. I think I can do that by treating the circuit as a voltage divider taking the ratio of the impedance at the input versus the impedance at the primary coil. No? Inductors effectively in parallel also increase the expected resonant frequency. If you have this, . L1 . +-----UUU--+------+------+ . | + | | | . ( Vsrc ) === C R 3 L2 . | - | 3 . | | | | . +----------+------+------+ . _|_ GND You might expect the resonant frequency is L2 + C, but it's actually (L1 || L2) = Leq. If L1 is not substantially larger than L2, the resonant frequency will be pulled higher. I see, L1 and L2 are in parallel because the impedance of Vsrc is very low. That is not the circuit I am simulating however. The loop of the antenna and the loop of the inductor are in series along with the primary capacitor. I'm not sure what the resistor is intended to represent, perhaps transformer losses? The resistance of L1 was added to the simulation model along with the resistance of the secondary coil which you have not shown... I think. It seems to me you have left out the tuning capacitor on the primary. Incidentally, don't forget to include loss components. I didn't see any explict R on the schematic. I didn't check if you set the LTSpice default parasitic ESR (cap), or DCR or EPR (coil) on the components. Besides parasitic losses, your signal is going *somewhere*, and that "where" consumes power! The actual transmitter is most certainly not a perfect current source inductor, nor is the receiver lossless. This simulation has no expression for radiation in any direction that's not directly between the two antennas: if all the power transmitted by the current source is reflected back, even though it's through a 0.1% coupling coefficient, it has to go somewhere. If it's coming back out the antenna, and it's not being burned in the "transformer", it's coming back into the transmitter. This is at odds with reality, where a 100% reflective antenna doesn't magically smoke a distant transmitter, it simply reflects 99.9% back into space. The transmitter hardly knows. Interesting point. My primary goal with this is to simulate the resonance of the tuning so I can understand how to best tune the circuit. In many of the simulations I run the Q ends up being high enough that a very small drift in the parasitic capacitance on the secondary detunes the antenna and drops the signal level. It sounds like there are other losses that will bring the Q much lower. I would also like to have some idea of the signal strength to expect. My understanding is that the radiation resistance of loop antennas is pretty low. So not much energy will be radiated out. No? You make it sound as if in the simulation, even with a small coupling coefficient all the energy from antenna inductor will still couple back into the transmitter inductor regardless of the K value. Do I misunderstand you? It seems to result in the opposite, minimizing this back coupling. Or are you saying that the simulation needs to simulate the radiation resistance to show radiated losses? In this example, if you set R very large, you'll see ever more voltage on the output, and ever more current draw from Vsrc. You can mitigate this by increasing L1 still further, but the point is, if the source and load (R) aren't matched in some fashion, the power will reflect back to the transmitter and cause problems (in this case, power reflected back in-phase causes excessive current draw; in the CCS case, reflected power in-phase causes minimal voltage generation and little power transmission). Power is always coming and going somewhere, and if you happen to forget this fact, it'll reflect back and zap you in the butt sooner or later! Tim Actually, my goal was to build the receiver and I realized that my design would require the largest signal I could get from the antenna. I never realized I would end up having to learn quite so much about antenna design. I've been planning to create a PCB with lots of options so I can test a number of configurations. Nothing about the simulation makes me doubt the utility of this idea. One thing that continues to bug me is that nothing I have seen gives me a hint on how to factor in the distributed capacitance of the antenna shield. I am using RG6 with 16 pF/Ft and likely will end up with 100 foot of coax total. At some point I'll just have to make some measurements and see what the real world does. -- Rick |
Antenna Simulation in LTspice
On 3/1/2013 11:43 AM, legg wrote:
On Wed, 27 Feb 2013 22:07:51 -0500, wrote: snip There is a resonance near the frequency I would expect, but it is not so close actually. I can't figure why it is about 5% off. There is a second resonance fairly high up that I can't figure at all. None of the component values seem to combine appropriately to produce this peak. snip Pulling out the old reactance paper, there are a couple of expected interactions using the values present: Around 50KHz (89.42uH+48uH) with 50.42nF (L3+L1) with C1 Around 290KHz (89.42uH+48uH) with 6.25nF (L3+L1) with C2*N^2 Around 360KHz 48uH with 6.25nF L1 with C2*N^2 nL1/nL2=N=25 The mid-resonance is a dip or rejection. What's the issue? 290 kHz matches the calculations you just gave. But 290 kHz is the null (or dip as you call it) from C2 and L2 (or L1 and C2 reflected with N^2). I thought I wasn't getting the 60 kHz resonance, but I was mistakenly adding the two capacitances together. So that is closer. Using L3+L1 with C1 I get 60.46 kHz while it is measured at 60 dead on in simulation. That's nearly a 1% error. I solved the equations finally. I found some info on the impedance of series and parallel circuits. With that info I wrote the equation for the ratio of Vout/Vin and found the roots. Turns out it is not so bad. The equation is a fourth order, but it has no x^3 or x^1 terms and so is actually a quadratic of x^2. Solving the quadratic gives the exact figures for 60 kHz and 393 kHz peaks. Since this is from taking the square root of x^2, there are also solutions at the negative values... duh! Reflecting C2 through the transformer to create C2', the two nulls I found can be calculated by the resonance of L1 and C2' (290 kHz null on C1) or L1 with C1 and C2' (96500 Hz null on L3). -- Rick |
Antenna Simulation in LTspice
On 3/1/2013 8:53 PM, Tim Williams wrote:
You'll be much better off simply using the conventional radio approach than trying to simulate everything, especially when circuit equivalents are nebulous like this. I don't know what you mean by the "conventional radio approach". After all, if you can't quite tell what it *should* look like, how would you know if you could implement your model once you've found a satisfactory result? I was simulating a specific circuit for a specific purpose. I got the answer I was looking for. What kind of antenna are you looking at, loop? The first thing to know about a loop is, if it's a very small loop (I'm guessing, at this frequency, it is), its radiation resistance is very low, meaning, you can treat it as a nearly pure inductance (Q 10 I think is typical), and its bandwidth (even with a matched load) will be correspondingly narrow. Yes, I plan to use a shielded loop. I have found some contradictory info on the effectiveness of the "shield". One reference seems to have measurements that show it is primarily E-field coupled in the longer distance portion of the near-field. I am aware of the low radiation resistance and have not included that factor in my simulation. The Q of just the antenna loop is around 100 as calculated from the ratio of reactance to resistance. The nature of the incoming signal could be modeled as a voltage or current source; how doesn't really matter, because it isn't really either, it's a power source that couples in. Again, you don't have voltage without current and vice versa, it's all about power flow, and the matching that allows the power to flow. A friend in a loop antenna Yahoo group suggested the use of the transformer coupling with a low k to model the signal reception. Since the loop is inductive, your first priority is to resonate it with a capacitor at the desired frequency. This will require a very precise value, and even for a single frequency, may require a variable capacitor to account for manufacturing tolerances. In the AM BCB, a Q of 10 gets you 50-160kHz bandwidth, so you only get a few channels for any given tuning position. And if the Q is higher, you get even fewer. Yes, that is loop antenna 101 I think. It was when I added a coupling transformer with 100:1 turns ratio that I was told I needed to consider the parasitics. I have found it is not useful to go much above 25 or 33:1 on the turns ratio. I am receiving a single frequency, 60 kHz. There is no need for a wide bandwidth. Ultimately, I prefer a Q of 100 for the higher gain. If it gets too high, the off tuning by variations (drift) in the parasitic capacitance affects the antenna gain appreciably. Now that you've got a high Q resonant tank, you can do two things: couple into the voltage across the capacitor, or the current through the inductor. You need only a small fraction of either, because the Q is still going to be large. This can be arranged with a voltage divider (usually the capacitor is split into a huge hunk and a small variable part, e.g., 300pF variable + 10nF, output from across the 10nF), a transformer (a potential transformer across the cap, or a current transformer in series with the inductor), an inductive pickup (the big loop carries lots of volts, but you only need a few, so a much smaller loop can be placed inside the big loop), an impractically large inductor (like in my example circuit, which models radiation resistance as a parallel equivalent), etc. Whatever the case, you need to match transmission line impedance (e.g., 50 ohms) to radiation resistance (whichever series or parallel equivalent you have). Transmission line? What transmission line? The antenna is directly connected to the receiver which has a very high input impedance. Why do I need to consider radiation resistance? I have not read that anywhere. Once you get the signal into a transmission line, with a reasonable match (Z ~= Z_line, or alternately, SWR ~= 1), you can do whatever you want with it. Put it into an amplifier (don't forget to match it, too), etc. Yes, you're going to have funny behavior at other frequencies, and if you're concerned about those frequencies, you'll have to choose the coupling circuit and adjustable (or selectable) components accordingly. But for the most part, you completely ignore any frequency that you aren't tuning for, usually enforcing that concept by inserting filters to reject any stragglers. Example: suppose you have a loop of 5uH and need to tune it to 500kHz. It has a reactance of 15.7 ohms. Suppose further it has Q = 20. The ESR (not counting DCR and skin effect) is X_L / Q, or 0.78 ohms; alternately, the EPR is X_L * Q, or 314 ohms. The capacitor required is 20.3nF. If we use a current transformer to match to a 50 ohm line, it needs an impedance ratio of 1:64, or a turns ratio of 1:8. If we use a voltage transformer, it's of course 8:1. (A capacitor divider is unsuitable for resonant impedances less than line impedance, since it can only divide the impedance down. If the inductance were a lot larger, it could be used.) To a rough approximation, a smaller inductive loop, of 1/8 diameter of the larger, I think, would also work. I'm not familiar with the concept of voltage transformer vs. current transformer. How do you mean that? How did you get the 1:64 impedance ratio and the 1:8 turns ratio? I don't follow that. Are you saying the line impedance should match the ESR? Why exactly would it need to match the ESR? -- Rick |
Antenna Simulation in LTspice
"rickman" wrote in message
... Yes, I plan to use a shielded loop. I have found some contradictory info on the effectiveness of the "shield". One reference seems to have measurements that show it is primarily E-field coupled in the longer distance portion of the near-field. I trust this resource: http://vk1od.net/antenna/shieldedloop/ He's got gobs of analytical articles. Yes, that is loop antenna 101 I think. It was when I added a coupling transformer with 100:1 turns ratio that I was told I needed to consider the parasitics. I have found it is not useful to go much above 25 or 33:1 on the turns ratio. I am receiving a single frequency, 60 kHz. There is no need for a wide bandwidth. Ultimately, I prefer a Q of 100 for the higher gain. If it gets too high, the off tuning by variations (drift) in the parasitic capacitance affects the antenna gain appreciably. High Q isn't the goal, high radiation resistance is -- the bigger the loop, the better it couples with free space, until it's a wave length around. You can go ahead and make a teeny coil out of polished silver litz wire, and push the Q up into the hundreds, but all you'll see is internal resistance, hardly anything attributable to actual radiation. Since the losses dominate over radiation, it makes a crappy antenna. But you know that from looking at it -- it's a tiny lump, of course it's not going to see the outside world. It is true, however, that a small coil, with low losses, will have low noise. AM radios rely on this, which is how they get away with tiny hunks of ferrite for picking up radio. Of course, it doesn't hurt that AM stations are 50kW or so, to push over atmospheric noise. Transmission line? What transmission line? The antenna is directly connected to the receiver which has a very high input impedance. Why do I need to consider radiation resistance? I have not read that anywhere. Ok, then you can merge the matching transformer, transmission line and receiver input transformer into one -- an even larger stepup into whatever impedance it's looking at (what's "very high", kohms? Mohms?) will get you that much more SNR. I'm not familiar with the concept of voltage transformer vs. current transformer. How do you mean that? Current transformer measures current (its winding is in series), potential transformer measures voltage (in parallel). How did you get the 1:64 impedance ratio and the 1:8 turns ratio? I don't follow that. Are you saying the line impedance should match the ESR? Why exactly would it need to match the ESR? ESR (and Q) measured on the coil corresponds to radiation resistance (series equivalent) *plus* internal losses (also series equivalent). You can't separate the two components, so you can only get the best power match by the good old impedance theorem. ~1:64 is 50 ohm / 0.78 ohm, and N2/N1 = sqrt(Z2/Z1), or 8:1 turns ratio. Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com |
Antenna Simulation in LTspice
On 6.3.13 9:00 , rickman wrote:
Since the loop is inductive, your first priority is to resonate it with a capacitor at the desired frequency. This will require a very precise value, and even for a single frequency, may require a variable capacitor to account for manufacturing tolerances. In the AM BCB, a Q of 10 gets you 50-160kHz bandwidth, so you only get a few channels for any given tuning position. And if the Q is higher, you get even fewer. Yes, that is loop antenna 101 I think. It was when I added a coupling transformer with 100:1 turns ratio that I was told I needed to consider the parasitics. I have found it is not useful to go much above 25 or 33:1 on the turns ratio. I am receiving a single frequency, 60 kHz. There is no need for a wide bandwidth. Ultimately, I prefer a Q of 100 for the higher gain. If it gets too high, the off tuning by variations (drift) in the parasitic capacitance affects the antenna gain appreciably. Please note that high Q will destroy the modulation sidebands on the signal you're listening to. In aviation, there are non-directional beacons which are transmitting in a frequency around 300 kHz (1 km wavelength). The antennas cannot obviously be of efficient length (250 m / 800 ft), so they are short (20 m / 70 ft) force-tuned to the transmitting frequency. This creates so high Q that the identification modulation sidebands for the customary 1050 Hz audio do not fit in, and the ID is modulated using 400 Hz audio. -- Tauno Voipio, avionics engineer (also OH2UG) |
Antenna Simulation in LTspice
On 3/7/2013 2:15 AM, Tauno Voipio wrote:
On 6.3.13 9:00 , rickman wrote: Since the loop is inductive, your first priority is to resonate it with a capacitor at the desired frequency. This will require a very precise value, and even for a single frequency, may require a variable capacitor to account for manufacturing tolerances. In the AM BCB, a Q of 10 gets you 50-160kHz bandwidth, so you only get a few channels for any given tuning position. And if the Q is higher, you get even fewer. Yes, that is loop antenna 101 I think. It was when I added a coupling transformer with 100:1 turns ratio that I was told I needed to consider the parasitics. I have found it is not useful to go much above 25 or 33:1 on the turns ratio. I am receiving a single frequency, 60 kHz. There is no need for a wide bandwidth. Ultimately, I prefer a Q of 100 for the higher gain. If it gets too high, the off tuning by variations (drift) in the parasitic capacitance affects the antenna gain appreciably. Please note that high Q will destroy the modulation sidebands on the signal you're listening to. I appreciate the advice from everyone, but much of it is not in the proper context and way off target. "High" Q is how high? Where are the modulation sidebands? My point is that I have already considered this. The modulation sidebands of this signal are on the order of low 10's of Hz. This signal is modulated at a 1 bit per second rate. I will be demodulating a 30 Hz sample rate. So a bandwidth of 100 Hz is plenty which corresponds to a Q of around 500. I said I was looking for a Q over 100, maybe I should have said a Q of a bit over 100. By the time it gets to 300 it is to peaky to hold a tune setting. That is the problem I am concerned with. In aviation, there are non-directional beacons which are transmitting in a frequency around 300 kHz (1 km wavelength). The antennas cannot obviously be of efficient length (250 m / 800 ft), so they are short (20 m / 70 ft) force-tuned to the transmitting frequency. This creates so high Q that the identification modulation sidebands for the customary 1050 Hz audio do not fit in, and the ID is modulated using 400 Hz audio. Ok, but that is nothing like my application, receiving WWVB. -- Rick |
Antenna Simulation in LTspice
On 3/6/2013 8:13 PM, Tim Williams wrote:
wrote in message ... Yes, I plan to use a shielded loop. I have found some contradictory info on the effectiveness of the "shield". One reference seems to have measurements that show it is primarily E-field coupled in the longer distance portion of the near-field. I trust this resource: http://vk1od.net/antenna/shieldedloop/ He's got gobs of analytical articles. Yes, I've seen this page. Thanks. Yes, that is loop antenna 101 I think. It was when I added a coupling transformer with 100:1 turns ratio that I was told I needed to consider the parasitics. I have found it is not useful to go much above 25 or 33:1 on the turns ratio. I am receiving a single frequency, 60 kHz. There is no need for a wide bandwidth. Ultimately, I prefer a Q of 100 for the higher gain. If it gets too high, the off tuning by variations (drift) in the parasitic capacitance affects the antenna gain appreciably. High Q isn't the goal, high radiation resistance is -- the bigger the loop, the better it couples with free space, until it's a wave length around. I'm not clear on why you keep referring to radiation resistance for a receiving antenna. Does this result in a larger received signal? I am concerned with maximizing the voltage at the input to the receiver. You can go ahead and make a teeny coil out of polished silver litz wire, and push the Q up into the hundreds, but all you'll see is internal resistance, hardly anything attributable to actual radiation. Since the losses dominate over radiation, it makes a crappy antenna. But you know that from looking at it -- it's a tiny lump, of course it's not going to see the outside world. I have no idea why you are talking about Litz wire and tiny coils. I never said I was looking to maximize the Q. I said I wanted a Q of over 100. I should have said, slightly over 100. A higher Q clearly does increase the voltage on the input in my simulations. Is there something wrong with my simulations? It is true, however, that a small coil, with low losses, will have low noise. AM radios rely on this, which is how they get away with tiny hunks of ferrite for picking up radio. Of course, it doesn't hurt that AM stations are 50kW or so, to push over atmospheric noise. Transmission line? What transmission line? The antenna is directly connected to the receiver which has a very high input impedance. Why do I need to consider radiation resistance? I have not read that anywhere. Ok, then you can merge the matching transformer, transmission line and receiver input transformer into one -- an even larger stepup into whatever impedance it's looking at (what's "very high", kohms? Mohms?) will get you that much more SNR. Yes, a higher stepup ratio gets larger signal up to a point. That point is determined by the parasitic capacitance of the receiver input. That capacitance is reflected back through the transformer and affects the antenna tuning. In my simulations it creates a filter with two resonances. I'm not familiar with the concept of voltage transformer vs. current transformer. How do you mean that? Current transformer measures current (its winding is in series), potential transformer measures voltage (in parallel). Series and parallel with what? I'm not following this. I have trouble with series and parallel resonance, but I'm starting to get the concept. Sometimes it is hard to tell how a circuit is being stimulated. How did you get the 1:64 impedance ratio and the 1:8 turns ratio? I don't follow that. Are you saying the line impedance should match the ESR? Why exactly would it need to match the ESR? ESR (and Q) measured on the coil corresponds to radiation resistance (series equivalent) *plus* internal losses (also series equivalent). You can't separate the two components, so you can only get the best power match by the good old impedance theorem. Internal losses of what? How do you determine the internal losses? ~1:64 is 50 ohm / 0.78 ohm, and N2/N1 = sqrt(Z2/Z1), or 8:1 turns ratio. Ok, so you were matching the hypothetical ESR to the hypothetical line impedance. -- Rick |
Antenna Simulation in LTspice
On 7.3.13 4:30 , rickman wrote:
On 3/7/2013 2:15 AM, Tauno Voipio wrote: On 6.3.13 9:00 , rickman wrote: Since the loop is inductive, your first priority is to resonate it with a capacitor at the desired frequency. This will require a very precise value, and even for a single frequency, may require a variable capacitor to account for manufacturing tolerances. In the AM BCB, a Q of 10 gets you 50-160kHz bandwidth, so you only get a few channels for any given tuning position. And if the Q is higher, you get even fewer. Yes, that is loop antenna 101 I think. It was when I added a coupling transformer with 100:1 turns ratio that I was told I needed to consider the parasitics. I have found it is not useful to go much above 25 or 33:1 on the turns ratio. I am receiving a single frequency, 60 kHz. There is no need for a wide bandwidth. Ultimately, I prefer a Q of 100 for the higher gain. If it gets too high, the off tuning by variations (drift) in the parasitic capacitance affects the antenna gain appreciably. Please note that high Q will destroy the modulation sidebands on the signal you're listening to. I appreciate the advice from everyone, but much of it is not in the proper context and way off target. "High" Q is how high? Where are the modulation sidebands? My point is that I have already considered this. The modulation sidebands of this signal are on the order of low 10's of Hz. This signal is modulated at a 1 bit per second rate. I will be demodulating a 30 Hz sample rate. So a bandwidth of 100 Hz is plenty which corresponds to a Q of around 500. I said I was looking for a Q over 100, maybe I should have said a Q of a bit over 100. By the time it gets to 300 it is to peaky to hold a tune setting. That is the problem I am concerned with. In aviation, there are non-directional beacons which are transmitting in a frequency around 300 kHz (1 km wavelength). The antennas cannot obviously be of efficient length (250 m / 800 ft), so they are short (20 m / 70 ft) force-tuned to the transmitting frequency. This creates so high Q that the identification modulation sidebands for the customary 1050 Hz audio do not fit in, and the ID is modulated using 400 Hz audio. Ok, but that is nothing like my application, receiving WWVB. I'd still be wary of high Q. The antenna is, by definition, in close interaction with its surroundings, and a high-Q thing is quickly detuned. At those low frequencies, the atmospheric and other outside noise is far larger than the internal noise of an amplifier, so in my opinion, the way to go is a loop tuned to 60 kHz with as low Q as easily comes without extra attenuation and a good pre-amplifier. The preamp can then contain a tuned interstage tank for interference suppression. -- Tauno Voipio |
Antenna Simulation in LTspice
On 3/7/2013 12:55 PM, Tauno Voipio wrote:
On 7.3.13 4:30 , rickman wrote: Ok, but that is nothing like my application, receiving WWVB. I'd still be wary of high Q. The antenna is, by definition, in close interaction with its surroundings, and a high-Q thing is quickly detuned. At those low frequencies, the atmospheric and other outside noise is far larger than the internal noise of an amplifier, so in my opinion, the way to go is a loop tuned to 60 kHz with as low Q as easily comes without extra attenuation and a good pre-amplifier. The preamp can then contain a tuned interstage tank for interference suppression. I understand. But this is intended to be *very* low power and I haven't found an amp I can use that is in the low double digits uW power consumption range. I plan to use no amp and go straight to digital. -- Rick |
Antenna Simulation in LTspice
On Thu, 07 Mar 2013 13:11:03 -0500, rickman wrote:
I understand. But this is intended to be *very* low power and I haven't found an amp I can use that is in the low double digits uW power consumption range. I plan to use no amp and go straight to digital. I don't think that's possible. Unless your input A/D converter can operate in the microvolt region, it's going to have a difficult time dealing with the low signal levels. Fortunately, WWVB is on-off keying with no amplitude component, so there's no incentive to add an AGC controlled input amplifier in order to maximize the A/D converters dynamic range. Still, you need to work with something more than a few bits above the noise level. Incidentally, after midnight, you WWVB delivers about 100 uV/meter or more to continental US. http://tf.nist.gov/tf-cgi/wwvbmonitor_e.cgi (Java required) I've seen it strong enough that I can see the waveform on an oscilloscope after a 60Khz passive filter. As for bandwidth, the code is sent at 1 baud (1 bit/sec) which produces about a 2Hz occupied bandwidth. Therefore, the maximum Q of the antenna would need to be: 60Khz/ 2Hz = 30,000 before the antenna bandwidth becomes a problem. Incidentally, while Googling away merrily, I found this on SPICE models for a loop antenna. It's not quite in your xformer format, but it might be useful: http://sidstation.loudet.org/antenna-theory-en.xhtml I won't pretend to understand what the author is doing until I read it more carefully. Incidentally, I used a WWVB code simulator driving a signal generator to test my receiver: http://www.leapsecond.com/notes/wwvb2.htm If you're seriously into this, I suggest asking questions on the time-nuts mailing list: https://www.febo.com/mailman/listinfo/time-nuts -- Jeff Liebermann 150 Felker St #D http://www.LearnByDestroying.com Santa Cruz CA 95060 http://802.11junk.com Skype: JeffLiebermann AE6KS 831-336-2558 |
Antenna Simulation in LTspice
"rickman" wrote in message
... High Q isn't the goal, high radiation resistance is -- the bigger the loop, the better it couples with free space, until it's a wave length around. I'm not clear on why you keep referring to radiation resistance for a receiving antenna. Does this result in a larger received signal? I am concerned with maximizing the voltage at the input to the receiver. You're also not concerned about that -- you're concerned about maximizing SNR at the receiver. A Q of a million will get you gobs of "gain", but if it doesn't couple into free space, it's only the thermal noise of the loss generating that signal. An antenna with high (expressed as ESR) radiation resistance might have a modest Q, but gives far better SNR because it couples to free space. Raw volts don't matter, you can always throw more amplifiers at it (as long as they don't corrupt the SNR also!). Yes, a higher stepup ratio gets larger signal up to a point. That point is determined by the parasitic capacitance of the receiver input. That capacitance is reflected back through the transformer and affects the antenna tuning. In my simulations it creates a filter with two resonances. Oooh, capacitance! I like capacitance. Capacitance is easy to cancel...inductors are good at that. :) What's a nearby inductor working against that capacitance? The current transformer in your simulation, if its inductance can be controlled, would be an excellent candidate. The circuit effectively becomes a double tuned interstage transformer, like, http://www.jrmagnetics.com/rf/doubtune/doubccl_c.php This is two resonators coupled with a cap, but any coupling method will do. Capacitive, magnetic (putting the coils end-to-end) or electromagnetic (coils side-by-side) coupling does equally well; normal arrangements have them all in phase, so in practice, unshielded coils will need smaller coupling capacitance than designed, etc. If you line up that 10p resonance with the operating frequency, you should get gobs more gain. In fact, because the reactances cancel, the driven impedance will be much higher than you were expecting, and so will the gain. The CT might go from, say, 1:8 up to, who knows, 1:20? 1:100? The bandwidth of that coupling (not necessarily of the antenna itself, so they should be similar bandwidths) is determined by the coupling coefficient (in the coupled-inductors case, simply k) and Q of the components. If your receiver datasheet specifies an equivalent input circuit, you might be able to estimate the equivalent loss and optimize gain. Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com |
Antenna Simulation in LTspice
On 3/7/2013 5:14 PM, Tim Williams wrote:
wrote in message ... High Q isn't the goal, high radiation resistance is -- the bigger the loop, the better it couples with free space, until it's a wave length around. I'm not clear on why you keep referring to radiation resistance for a receiving antenna. Does this result in a larger received signal? I am concerned with maximizing the voltage at the input to the receiver. You're also not concerned about that -- you're concerned about maximizing SNR at the receiver. SNR would be good, but I am concerned with maximizing the signal actually. A Q of a million will get you gobs of "gain", but if it doesn't couple into free space, it's only the thermal noise of the loss generating that signal. I think you aren't reading what I am writing. I said I wanted a Q over 100, not 1 million. I don't get why you keep talking in hyperbole. What you are describing is not even a tradeoff between signal strength and SNR. If there is no coupling, there is no signal. An antenna with high (expressed as ESR) radiation resistance might have a modest Q, but gives far better SNR because it couples to free space. I have not found anything to indicate this produces a better receive antenna. I have a formula for the effective height of a loop antenna which is what determines the received signal strength at the antenna. It does not calculate the radiation resistance, it uses the coil parameters and the wire resistance. Is that a wrong formula? Raw volts don't matter, you can always throw more amplifiers at it (as long as they don't corrupt the SNR also!). Maybe you didn't read my other posts. I am not using an amplifier. I am running the antenna and coupler output directly into a digital input. Yes, a higher stepup ratio gets larger signal up to a point. That point is determined by the parasitic capacitance of the receiver input. That capacitance is reflected back through the transformer and affects the antenna tuning. In my simulations it creates a filter with two resonances. Oooh, capacitance! I like capacitance. Capacitance is easy to cancel...inductors are good at that. :) What's a nearby inductor working against that capacitance? The current transformer in your simulation, if its inductance can be controlled, would be an excellent candidate. The circuit effectively becomes a double tuned interstage transformer, like, http://www.jrmagnetics.com/rf/doubtune/doubccl_c.php This is two resonators coupled with a cap, but any coupling method will do. Capacitive, magnetic (putting the coils end-to-end) or electromagnetic (coils side-by-side) coupling does equally well; normal arrangements have them all in phase, so in practice, unshielded coils will need smaller coupling capacitance than designed, etc. If you line up that 10p resonance with the operating frequency, you should get gobs more gain. In fact, because the reactances cancel, the driven impedance will be much higher than you were expecting, and so will the gain. The CT might go from, say, 1:8 up to, who knows, 1:20? 1:100? The bandwidth of that coupling (not necessarily of the antenna itself, so they should be similar bandwidths) is determined by the coupling coefficient (in the coupled-inductors case, simply k) and Q of the components. If your receiver datasheet specifies an equivalent input circuit, you might be able to estimate the equivalent loss and optimize gain. The receiver input is high impedance, approximately 10 MOhms with a low capacitance between the differential inputs of not more than 10 pF. Your description of what is happening is very terse and full of shortened terms that I don't understand. What do you mean "line up that 10p resonance with the operating frequency"? I assume you are referring to the 10 pF input capacitance. How does this get "lined up" with anything? When you talk about reactances canceling, that sounds a lot like a tuned circuit at resonance. That is what I *am* doing and where this thread started. One problem with that is the lack of precision or stability of the parasitic capacitance. Any idea how to deal with that? Have you looked at the simulation data I had posted? I think you are describing exactly the circuit we are simulating which I believe is an accurate representation of the circuit I plan to build. Is that not correct? -- Rick |
Antenna Simulation in LTspice
On Thu, 07 Mar 2013 09:50:11 -0500, rickman wrote:
On 3/6/2013 8:13 PM, Tim Williams wrote: wrote in message snip I'm not familiar with the concept of voltage transformer vs. current transformer. How do you mean that? Current transformer measures current (its winding is in series), potential transformer measures voltage (in parallel). Series and parallel with what? I'm not following this. snip An electric circuit consists of a source of power, a load, and something (like wires) connecting them. Transformers can be used if the source is providing alternating current. A voltage transformer is connected in parallel with the load so that the source, the transformer, and the load all see the same voltage. It can also be used to match a load to a source. A common example of a voltage transformer is the power transformer in a piece of equipment that changes the AC line voltage to whatever other voltages are required by the equipment. A current transformer, on the other hand, is connected in series with the load so that the source, load, and transformer all have the same current flowing through them. The most common use of a current transformer is to measure the current flowing into a load. A clamp-on ammeter is a common example. Historical examples of voltage and current transformers are the "picture tube brighteners" that were commonly used in TV sets to prolong the useful life of the CRT. There were two types, parallel and series. The parallel types were used in transformer operated TVs and consisted of a step-up transformer to raise the heater voltage of the CRT above normal to increase emission. The series type was used in sets with the tube heaters in series and consisted of a step-down transformer that raised the heater current above normal. Of course, raising either the voltage or the current also raised the other. These were, respectively, voltage and current transformers. A loop antenna is a distributed source with the voltage being generated along the length of the wire and also having a magnetic field so that it can be used as part of a transformer. This blurs the distinction between a current and voltage transformer. -- Jim Mueller To get my real email address, replace wrongname with dadoheadman. Then replace nospam with fastmail. Lastly, replace com with us. |
Antenna Simulation in LTspice
On Thu, 07 Mar 2013 11:39:45 -0800, Jeff Liebermann
wrote: Fortunately, WWVB is on-off keying with no amplitude component, so there's no incentive to add an AGC controlled input amplifier in order to maximize the A/D converters dynamic range. Oops. WWVB does have an amplitude component and is not quite on-off keying (OOK). There's a -17dB drop in RF signal level at the beginning of each 1 second marker pulse. It was -10dB prior to 2005. http://en.wikipedia.org/wiki/WWVB#Modulation_depth -- Jeff Liebermann 150 Felker St #D http://www.LearnByDestroying.com Santa Cruz CA 95060 http://802.11junk.com Skype: JeffLiebermann AE6KS 831-336-2558 |
Antenna Simulation in LTspice
On 3/7/2013 2:39 PM, Jeff Liebermann wrote:
On Thu, 07 Mar 2013 13:11:03 -0500, wrote: I understand. But this is intended to be *very* low power and I haven't found an amp I can use that is in the low double digits uW power consumption range. I plan to use no amp and go straight to digital. I don't think that's possible. Unless your input A/D converter can operate in the microvolt region, it's going to have a difficult time dealing with the low signal levels. Fortunately, WWVB is on-off keying with no amplitude component, so there's no incentive to add an AGC controlled input amplifier in order to maximize the A/D converters dynamic range. Still, you need to work with something more than a few bits above the noise level. Incidentally, after midnight, you WWVB delivers about 100 uV/meter or more to continental US. http://tf.nist.gov/tf-cgi/wwvbmonitor_e.cgi (Java required) I've seen it strong enough that I can see the waveform on an oscilloscope after a 60Khz passive filter. Yes, I have done my homework on the WWVB signal. I am at the fringe of the 100 uV/m contour. I would very much like to see the signal on an oscilloscope when I test this. They have a receiver not far from here in Gaithersburg, MD and the signal is often strong during the day. So much so that I don't follow why they say there is this day/night signal strength fluctuation. It looks much more random to me. The WWVB signal is not truly on-off keying. I believe they use a 10 dB modulation factor for the AM signal. This is close to on-off I agree. But they also phase modulate the signal and I will be demodulating both to see which one works best in my design. The ADC in my design is truly one bit. It is an LVDS input on an FPGA. I looked at delta-sigma (or is it sigma-delta? ;) conversion and got code from the chip vendor for a simplistic implementation. I don't think I have the power budget for that and am using a much simpler 1 bit ADC at 4x the carrier rate. The bit stream is multiplied by quadrature carriers at 60 kHz and each stream summed for 1/30 of a second to implement what can be considered a DFT bin, a decimated FIR filter or a decimated down conversion; take your pick, they are all mathematically the same in this case because the sampling is synchronous to the carrier (or very close to synchronous). What comes out the other end of this processing gains nearly 40 dB in SNR. My simulations show a recoverable signal when it is more than 20 dB below the noise. Of course, I have not tested this yet on a real signal. I want to run some tests on the antenna and coupling transformer to verify the simulation. Then I will start working with the FPGA to see if I can make the LVDS input do what I want. I have ideas on how to bend digital circuits to do my bidding. This LVDS input is why I want as large a signal as possible from the antenna. With the high impedance input on the chip I should be able to boost the signal pretty well with just passive devices and signal processing. The loop antenna is rather large. I would like to end up with something smaller. Once I get this working with a shielded loop antenna I will check out the ferrite core antennas. My understanding is that they don't produce as much signal. As for bandwidth, the code is sent at 1 baud (1 bit/sec) which produces about a 2Hz occupied bandwidth. Therefore, the maximum Q of the antenna would need to be: 60Khz/ 2Hz = 30,000 before the antenna bandwidth becomes a problem. I'm not sure how you came up with 2 Hz for the bandwidth. In this case the bandwidth is not just twice the bit rate. I believe the stated "system" bandwidth is around 5 Hz (from a 1995 paper prior to addition of the phase modulation). Regardless, I am sampling at 30 Hz and if I expect to see significant changes in phase or amplitude within one sample time, I need an appropriate bandwidth. Even so, that is not the limiting factor. The limiting factor is the difficulty in holding tune with drift in passive component values. The Q can be raised by increasing the turns ratio on the transformer, but it becomes so sensitive to the parasitic capacitance that the sensitivity drops 10 dB with a 1 pF change. Incidentally, while Googling away merrily, I found this on SPICE models for a loop antenna. It's not quite in your xformer format, but it might be useful: http://sidstation.loudet.org/antenna-theory-en.xhtml I won't pretend to understand what the author is doing until I read it more carefully. Thanks. I will take a look at that. Incidentally, I used a WWVB code simulator driving a signal generator to test my receiver: http://www.leapsecond.com/notes/wwvb2.htm I will be needing a time code simulator. I designed a commercial product that works with the IRIG-B time code which is similar. The functionality is not hard, it is just a matter of generating the data, encoding it into the modulation pattern, then impressing the carrier with the modulation. Working in an FPGA this sort of stuff is easy. The trouble is if you make the same mistake in both the generator and receiver they work just fine in simulation, but not with other equipment. lol I'll take a look at this link. If you're seriously into this, I suggest asking questions on the time-nuts mailing list: https://www.febo.com/mailman/listinfo/time-nuts I might look into that. Certainly it can't hurt to get more input. -- Rick |
Antenna Simulation in LTspice
On 3/7/2013 10:17 PM, Jim Mueller wrote:
On Thu, 07 Mar 2013 09:50:11 -0500, rickman wrote: On 3/6/2013 8:13 PM, Tim Williams wrote: wrote in message snip I'm not familiar with the concept of voltage transformer vs. current transformer. How do you mean that? Current transformer measures current (its winding is in series), potential transformer measures voltage (in parallel). Series and parallel with what? I'm not following this. snip An electric circuit consists of a source of power, a load, and something (like wires) connecting them. Transformers can be used if the source is providing alternating current. A voltage transformer is connected in parallel with the load so that the source, the transformer, and the load all see the same voltage. It can also be used to match a load to a source. A common example of a voltage transformer is the power transformer in a piece of equipment that changes the AC line voltage to whatever other voltages are required by the equipment. A current transformer, on the other hand, is connected in series with the load so that the source, load, and transformer all have the same current flowing through them. The most common use of a current transformer is to measure the current flowing into a load. A clamp-on ammeter is a common example. Historical examples of voltage and current transformers are the "picture tube brighteners" that were commonly used in TV sets to prolong the useful life of the CRT. There were two types, parallel and series. The parallel types were used in transformer operated TVs and consisted of a step-up transformer to raise the heater voltage of the CRT above normal to increase emission. The series type was used in sets with the tube heaters in series and consisted of a step-down transformer that raised the heater current above normal. Of course, raising either the voltage or the current also raised the other. These were, respectively, voltage and current transformers. A loop antenna is a distributed source with the voltage being generated along the length of the wire and also having a magnetic field so that it can be used as part of a transformer. This blurs the distinction between a current and voltage transformer. Is this a current transformer or a voltage transformer? .--------. .--------. | | | | | C||C VAC C||C Load | C||C | | | | `--------' `--------' -- Rick |
Antenna Simulation in LTspice
"rickman" wrote in message
... A Q of a million will get you gobs of "gain", but if it doesn't couple into free space, it's only the thermal noise of the loss generating that signal. I think you aren't reading what I am writing. I said I wanted a Q over 100, not 1 million. I don't get why you keep talking in hyperbole. What you are describing is not even a tradeoff between signal strength and SNR. If there is no coupling, there is no signal. It may sound like hyperbole, but it's mathematically sound. The midpoint theorem, for example, guarantees that, between two points, you must've hit some point inbetween, somewhere, as long as the function is continuous. More usefully, functions arising in electronics are often one-to-one, so it's not only true that you are guaranteed midpoints, but you'll find them in order, too. If you aren't looking at the extreme cases, you aren't doing your job. Whatever's left inbetween can simply be interpolated! The point here being, an antenna which doesn't couple into free space obviously has a crappy SNR. The signal level can be anything, it doesn't matter. The signal need not be small, because internal losses generate thermal noise. With sufficient Q, you can push that thermal noise up to your receiver threshold (which you said is an ADC) and detect signal. It'll be bandlimited, ~60kHz noise, a useless signal, but present nonetheless. In general, antennas which do couple strongly to free space have low Qs. A 1/2 wave resonant dipole has a Q of only 1 or 2, so bothering to call it resonant is actually kind of weak. This is similarly true for a large loop, which of course would be highly impractical here. So there must be some middle case where SNR is reasonably unaffected, which will be the best choice antenna. Since atmospheric noise dominates, the antenna can stand to be pretty small. Raw volts don't matter, you can always throw more amplifiers at it (as long as they don't corrupt the SNR also!). Maybe you didn't read my other posts. I am not using an amplifier. I am running the antenna and coupler output directly into a digital input. You hadn't mentioned that before... The receiver input is high impedance, approximately 10 MOhms with a low capacitance between the differential inputs of not more than 10 pF. Any ESR? Example, the ATmega series 10 bit ADC specifies, I think, around 10pF + 10k ESR (somewhat depending on how many mux switches it's going through to get there). Your description of what is happening is very terse and full of shortened terms that I don't understand. I could write a book on the subject to explore it in detail, but there are many available already, and there are too many holes in my knowledge to really be worth it, plus this is Usenet, you get what you pay for. I was hoping you'd Google in the blanks. What do you mean "line up that 10p resonance with the operating frequency"? I assume you are referring to the 10 pF input capacitance. How does this get "lined up" with anything? There's yet another theorem in networks that has to do with matching. A resonant tank's impedance varies wildly with frequency. But it will always be resistive at resonance. If you connect this to another network, which has a resistive input impedance at the same frequency, you don't care what the L and C are, it will simply work -- old fashioned resistor divider action! You *do* have to worry about L and C and reactance and bandwidth to solve for the frequency response and stuff, but you can at least approximate that with Q factor (i.e., how much loss is draining power out of the system). So if your ADC input is exactly 10p + 10M, you could resonate it with 0.7H (well...), which has a resonant impedance of 264k, and thus a reasonable Q of 38. (The real world typically bitchslaps the theorist at this point, as 0.7H chokes with 10pF parasitic capacitance and Q 38 at 60kHz don't exist.) If the capacitance's ESR is less than 6.9kohms (i.e., 264k / 38), it won't have significant effect. You can couple to this tank via parallel or series. If you did series, the input impedance would be 264k / 38, or 6.9k, not horrible; going from the 0.78 ohm loop to this in a single transformer requires a 1:100 CT, which works fine at 60kHz. (This CT would require high inductance, so as to avoid skewing results, but that's typical of a CT. An amorphous core CT would probably suffice. So at least that part is physically realizable.) Note the irony of coupling a current loop to a current loop, where in both cases, the CT looks like a small impedance relative to the loop it's within. That's simply how huge the impedance at the ADC is. Since all these resistances are matched, the power transfer theorem holds, and you're pushing as much voltage and power into the ADC as possible. The bandwidth is about 1.6kHz, so the thermal noise floor is around 5uV at the ADC. A received power of 1nW will generate 0.1V, which is probably a reasonable figure. The SNR of the receiver is limited by quantization noise for 14 bits, thermal for 14 bits. A 16 bit converter wouldn't be too expensive at this sample rate (note it's the analog sample-and-hold speed which limits direct conversion performance; a sigma-delta, running at 100Hz, with no S&H, won't see jack). When you talk about reactances canceling, that sounds a lot like a tuned circuit at resonance. That is what I *am* doing and where this thread started. One problem with that is the lack of precision or stability of the parasitic capacitance. Any idea how to deal with that? Considering theoretical 0.7H chokes aren't commercially available, you might swamp it with more C, which stabilizes the value, and requires less L to resonate. Rub: resonant impedance is lower, so the Q of the components must be higher in order to achieve the same performance. Even with a Q of 200, you still need over 0.25H, which is just as unlikely a combination. Well, if you really wanted to try, maybe a gapped ferrite-cored inductor could be made. Still, the only practical choice seems to be lower signal level. So ultimately, the question is, how little signal can you tolerate before you need an amplifier? How many bits of conversion, how much sample rate can you afford before a linear amplifier becomes cheaper on the power budget? Have you looked at the simulation data I had posted? I think you are describing exactly the circuit we are simulating which I believe is an accurate representation of the circuit I plan to build. Is that not correct? It's getting closer, but with adjustments (to the transformer inductance) to make the resonances line up (same frequencies). Plus whatever compromise you need to make on gain. Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com |
Antenna Simulation in LTspice
"rickman" wrote in message
... Is this a current transformer or a voltage transformer? .--------. .--------. | | | | | C||C VAC C||C Load | C||C | | | | `--------' `--------' Voltage. How about this? .--------. .--------. | | | | | C||C IAC C||C Load | C||C | | | | `--------' `--------' Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com |
Antenna Simulation in LTspice
On 3/8/2013 4:20 PM, Tim Williams wrote:
It may sound like hyperbole, but it's mathematically sound. The midpoint theorem, for example, guarantees that, between two points, you must've hit some point inbetween, somewhere, as long as the function is continuous. More usefully, functions arising in electronics are often one-to-one, so it's not only true that you are guaranteed midpoints, but you'll find them in order, too. If you aren't looking at the extreme cases, you aren't doing your job. Whatever's left inbetween can simply be interpolated! The point here being, an antenna which doesn't couple into free space obviously has a crappy SNR. The signal level can be anything, it doesn't matter. The signal need not be small, because internal losses generate thermal noise. With sufficient Q, you can push that thermal noise up to your receiver threshold (which you said is an ADC) and detect signal. It'll be bandlimited, ~60kHz noise, a useless signal, but present nonetheless. In general, antennas which do couple strongly to free space have low Qs. A 1/2 wave resonant dipole has a Q of only 1 or 2, so bothering to call it resonant is actually kind of weak. This is similarly true for a large loop, which of course would be highly impractical here. So there must be some middle case where SNR is reasonably unaffected, which will be the best choice antenna. I was warned a long time ago to be wary of people speaking in "glittering generalities". You seem to insist on using terms without giving a mathematical basis. How about if we use some math? V = (2 * pi * A * N * E * cos(theta)) / lambda V is the voltage on the antenna, A is the loop area, N is the number of turns, E is the field strength, theta is the rotation angle of the antenna and the transmitter (just consider this term to be 1, lamba is the wavelength (c/f) This is multiplied by the Q factor when resonated by a capacitor. So higher Q, higher signal. Where in here do you think I am having a problem? Since atmospheric noise dominates, the antenna can stand to be pretty small. You are making assumptions that don't hold true in my design. Raw volts don't matter, you can always throw more amplifiers at it (as long as they don't corrupt the SNR also!). Maybe you didn't read my other posts. I am not using an amplifier. I am running the antenna and coupler output directly into a digital input. You hadn't mentioned that before... You didn't ask. The receiver input is high impedance, approximately 10 MOhms with a low capacitance between the differential inputs of not more than 10 pF. Any ESR? Example, the ATmega series 10 bit ADC specifies, I think, around 10pF + 10k ESR (somewhat depending on how many mux switches it's going through to get there). FET input resistance. I will double check that though. Your description of what is happening is very terse and full of shortened terms that I don't understand. I could write a book on the subject to explore it in detail, but there are many available already, and there are too many holes in my knowledge to really be worth it, plus this is Usenet, you get what you pay for. I was hoping you'd Google in the blanks. I thought I was doing well, but you seem to be telling me I am making mistakes, but I can't figure out what they might be. What do you mean "line up that 10p resonance with the operating frequency"? I assume you are referring to the 10 pF input capacitance. How does this get "lined up" with anything? There's yet another theorem in networks that has to do with matching. A resonant tank's impedance varies wildly with frequency. But it will always be resistive at resonance. If you connect this to another network, which has a resistive input impedance at the same frequency, you don't care what the L and C are, it will simply work -- old fashioned resistor divider action! You *do* have to worry about L and C and reactance and bandwidth to solve for the frequency response and stuff, but you can at least approximate that with Q factor (i.e., how much loss is draining power out of the system). So if your ADC input is exactly 10p + 10M, you could resonate it with 0.7H (well...), which has a resonant impedance of 264k, and thus a reasonable Q of 38. (The real world typically bitchslaps the theorist at this point, as 0.7H chokes with10pF parasitic capacitance and Q 38 at 60kHz don't exist.) If the capacitance's ESR is less than 6.9kohms (i.e., 264k / 38), it won't have significant effect. You can couple to this tank via parallel or series. If you did series, the input impedance would be 264k / 38, or 6.9k, not horrible; going from the 0.78 ohm loop to this in a single transformer requires a 1:100 CT, which works fine at 60kHz. (This CT would require high inductance, so as to avoid skewing results, but that's typical of a CT. An amorphous core CT would probably suffice. So at least that part is physically realizable.) Note the irony of coupling a current loop to a current loop, where in both cases, the CT looks like a small impedance relative to the loop it's within. That's simply how huge the impedance at the ADC is. Since all these resistances are matched, the power transfer theorem holds, and you're pushing as much voltage and power into the ADC as possible. The bandwidth is about 1.6kHz, so the thermal noise floor is around 5uV at the ADC. A received power of 1nW will generate 0.1V, which is probably a reasonable figure. The SNR of the receiver is limited by quantization noise for14 bits, thermal for14 bits. A 16 bit converter wouldn't be too expensive at this sample rate (note it's the analog sample-and-hold speed which limits direct conversion performance; a sigma-delta, running at 100Hz, with no S&H, won't see jack). When you talk about reactances canceling, that sounds a lot like a tuned circuit at resonance. That is what I *am* doing and where this thread started. One problem with that is the lack of precision or stability of the parasitic capacitance. Any idea how to deal with that? Considering theoretical 0.7H chokes aren't commercially available, you might swamp it with more C, which stabilizes the value, and requires less L to resonate. Rub: resonant impedance is lower, so the Q of the components must be higher in order to achieve the same performance. Even with a Q of 200, you still need over 0.25H, which is just as unlikely a combination. Well, if you really wanted to try, maybe a gapped ferrite-cored inductor could be made. Still, the only practical choice seems to be lower signal level. So ultimately, the question is, how little signal can you tolerate before you need an amplifier? How many bits of conversion, how much sample rate can you afford before a linear amplifier becomes cheaper on the power budget? Have you looked at the simulation data I had posted? I think you are describing exactly the circuit we are simulating which I believe is an accurate representation of the circuit I plan to build. Is that not correct? It's getting closer, but with adjustments (to the transformer inductance) to make the resonances line up (same frequencies). Plus whatever compromise you need to make on gain. We are still having communications difficulties. You keep talking in terms I can't relate to. I don't need you to write a book, but I do need you to communicate clearly. I am using a 1 bit ADC. Don't assume that I am doing what you have done in the past. -- Rick |
Antenna Simulation in LTspice
On 3/8/2013 4:30 PM, Tim Williams wrote:
wrote in message ... Is this a current transformer or a voltage transformer? .--------. .--------. | | | | | C||C VAC C||C Load | C||C | | | | `--------' `--------' Voltage. How about this? .--------. .--------. | | | | | C||C IAC C||C Load | C||C | | | | `--------' `--------' Tim I have to say I don't follow the distinction. It is a transformer, no? -- Rick |
Antenna Simulation in LTspice
On Fri, 08 Mar 2013 13:44:36 -0500, rickman wrote:
I am at the fringe of the 100 uV/m contour. I would very much like to see the signal on an oscilloscope when I test this. I built a passive 60KHz bandpass filters out of a collection of ferrite cores from an old modem front end. I left it at a previous consulting job, but can resurrect the design if necessary. Incidentally, during my limited testing at home, I found that the biggest determent to decent reception was all the switching power supply noise found around the house. I finally ended up using a battery power oscilloscope http://802.11junk.com/jeffl/pics/drivel/slides/tek213.html a gel cell for powering the RF amp, and turning off the main power to the house. Then, I could sorta see a signal. They have a receiver not far from here in Gaithersburg, MD and the signal is often strong during the day. So much so that I don't follow why they say there is this day/night signal strength fluctuation. It looks much more random to me. http://tf.nist.gov/tf-cgi/wwvbgraph_e.cgi?5636103007 Very random. Compare the above graph with Santa Clara which looks less random: http://tf.nist.gov/tf-cgi/wwvbgraph_e.cgi?5636105007 On the east coast, besides a weak signal, you also have the potential for 60KHz interference from the UK: http://en.wikipedia.org/wiki/MSF_time_signal I had a 100KHz LORAN antenna on the roof of a former employer. The signal was just fine, until someone turned on the mercury vapor arc parking lot lamps at night. They were changed to low pressure sodium, which made testing possible at night. Incidentally, got any clue as to the vertical scale? My guess(tm) is 20 uv/meter signal strength per division, but I'm not sure. The WWVB signal is not truly on-off keying. I believe they use a 10 dB modulation factor for the AM signal. This is close to on-off I agree. It's now 17dB drop at the beginning of each UTC second. The change came in about 2008. But they also phase modulate the signal and I will be demodulating both to see which one works best in my design. The BPSK signal is much better at rejecting interference and digging the signal out of the noise. I don't know exactly how much, but I'm sure it's in a NIST publication somewhere. The ADC in my design is truly one bit. It is an LVDS input on an FPGA. I looked at delta-sigma (or is it sigma-delta? ;) It's delta-sigma. The loop antenna is rather large. I would like to end up with something smaller. Once I get this working with a shielded loop antenna I will check out the ferrite core antennas. My understanding is that they don't produce as much signal. Not exactly. Small loopsticks receive a proportional amount of noise. The ratio of signal to atmospheric noise remains roughly the same within a fixed bandwidth for any antenna. That's why tiny little loopsticks, inside "atomic time" wris****ches work. The small loopsticks also use the magnetic field instead of the electric field, which is why they can be made so small. http://en.wikipedia.org/wiki/Loop_antenna#Small_loops As for bandwidth, the code is sent at 1 baud (1 bit/sec) which produces about a 2Hz occupied bandwidth. Therefore, the maximum Q of the antenna would need to be: 60Khz/ 2Hz = 30,000 before the antenna bandwidth becomes a problem. I'm not sure how you came up with 2 Hz for the bandwidth. In this case the bandwidth is not just twice the bit rate. I believe the stated "system" bandwidth is around 5 Hz (from a 1995 paper prior to addition of the phase modulation). Ok, I made a bad guess(tm). Even at 5Hz BW, the maximum Q of 60KHz / 5Hz = 12,000 is not going to happen in a loop or loopstick antenna. Even so, that is not the limiting factor. The limiting factor is the difficulty in holding tune with drift in passive component values. Agreed. -- Jeff Liebermann 150 Felker St #D http://www.LearnByDestroying.com Santa Cruz CA 95060 http://802.11junk.com Skype: JeffLiebermann AE6KS 831-336-2558 |
Antenna Simulation in LTspice
On 3/10/2013 1:32 AM, Jeff Liebermann wrote:
On Fri, 08 Mar 2013 13:44:36 -0500, wrote: I am at the fringe of the 100 uV/m contour. I would very much like to see the signal on an oscilloscope when I test this. I built a passive 60KHz bandpass filters out of a collection of ferrite cores from an old modem front end. I left it at a previous consulting job, but can resurrect the design if necessary. Incidentally, during my limited testing at home, I found that the biggest determent to decent reception was all the switching power supply noise found around the house. I finally ended up using a battery power oscilloscope http://802.11junk.com/jeffl/pics/drivel/slides/tek213.html a gel cell for powering the RF amp, and turning off the main power to the house. Then, I could sorta see a signal. Holy crap! That's a lot of trouble to see a signal. By "see" I assume you mean on the scope. How large was the signal? The place where I am working currently is not very close to much and there isn't much in the house. I'm told the fridge is the biggest source of noise. We'll see how the CFL lamps do. Funny, last night my two RCC's both updated like they should. One is an analog clock and runs at 8x speed to get the hour ahead. In the fall it does this to go 11 hours ahead. Quite a sight! They both did the job, but my PC didn't update until it had been on for awhile, without being connected to the I'net. They have a receiver not far from here in Gaithersburg, MD and the signal is often strong during the day. So much so that I don't follow why they say there is this day/night signal strength fluctuation. It looks much more random to me. http://tf.nist.gov/tf-cgi/wwvbgraph_e.cgi?5636103007 Very random. Compare the above graph with Santa Clara which looks less random: http://tf.nist.gov/tf-cgi/wwvbgraph_e.cgi?5636105007 On the east coast, besides a weak signal, you also have the potential for 60KHz interference from the UK: http://en.wikipedia.org/wiki/MSF_time_signal Loop antennas have a null that can be steered toward the source of interference. I expect that will solve that problem... I had a 100KHz LORAN antenna on the roof of a former employer. The signal was just fine, until someone turned on the mercury vapor arc parking lot lamps at night. They were changed to low pressure sodium, which made testing possible at night. Incidentally, got any clue as to the vertical scale? My guess(tm) is 20 uv/meter signal strength per division, but I'm not sure. The WWVB signal is not truly on-off keying. I believe they use a 10 dB modulation factor for the AM signal. This is close to on-off I agree. It's now 17dB drop at the beginning of each UTC second. The change came in about 2008. But they also phase modulate the signal and I will be demodulating both to see which one works best in my design. The BPSK signal is much better at rejecting interference and digging the signal out of the noise. I don't know exactly how much, but I'm sure it's in a NIST publication somewhere. That's for an ideal receiver. I have my limitations and I have no idea how that will impact the reception. The ADC in my design is truly one bit. It is an LVDS input on an FPGA. I looked at delta-sigma (or is it sigma-delta? ;) It's delta-sigma. Actually I always say that with a smiley as it can be either. The loop antenna is rather large. I would like to end up with something smaller. Once I get this working with a shielded loop antenna I will check out the ferrite core antennas. My understanding is that they don't produce as much signal. Not exactly. Small loopsticks receive a proportional amount of noise. The ratio of signal to atmospheric noise remains roughly the same within a fixed bandwidth for any antenna. That's why tiny little loopsticks, inside "atomic time" wris****ches work. The small loopsticks also use the magnetic field instead of the electric field, which is why they can be made so small. http://en.wikipedia.org/wiki/Loop_antenna#Small_loops In my case I am not worried that the SNR isn't better, I just need a strong enough signal to drive the LVDS input. I will be providing feedback to eliminate any DC bias, but even that will only be so good. The input is claimed to have no hysteresis, but even a tiny amount can ruin this design. I will only know if this will work when I try it. As for bandwidth, the code is sent at 1 baud (1 bit/sec) which produces about a 2Hz occupied bandwidth. Therefore, the maximum Q of the antenna would need to be: 60Khz/ 2Hz = 30,000 before the antenna bandwidth becomes a problem. I'm not sure how you came up with 2 Hz for the bandwidth. In this case the bandwidth is not just twice the bit rate. I believe the stated "system" bandwidth is around 5 Hz (from a 1995 paper prior to addition of the phase modulation). Ok, I made a bad guess(tm). Even at 5Hz BW, the maximum Q of 60KHz / 5Hz = 12,000 is not going to happen in a loop or loopstick antenna. Even so, that is not the limiting factor. The limiting factor is the difficulty in holding tune with drift in passive component values. Agreed. -- Rick |
Antenna Simulation in LTspice
On Fri, 08 Mar 2013 16:51:57 -0500, rickman wrote:
On 3/8/2013 4:30 PM, Tim Williams wrote: wrote in message ... Is this a current transformer or a voltage transformer? .--------. .--------. | | | | | C||C VAC C||C Load | C||C | | | | `--------' `--------' Voltage. How about this? .--------. .--------. | | | | | C||C IAC C||C Load | C||C | | | | `--------' `--------' Tim I have to say I don't follow the distinction. It is a transformer, no? The second one is a current transformer. They both consist of coils around a magnetic core driving some kind of load. The difference is the source of power and that causes them to behave very differently as well as being constructed differently. Let's assume ideal components (a good place to start when learning a new concept). The voltage transformer is driven by a source that provides a constant voltage, no matter what the load. The transformer takes this voltage and converts it to some other voltage depending on the turns ratio; Vout = Vin * Ts / Tp. For example, if the primary has 100 turns and the secondary has 20 turns and the primary is supplied with 50 volts, the secondary will provide 10 volts. As the secondary load changes, this voltage remains the same but the current changes. If the secondary is open-circuited, the voltage still stays the same. If the secondary is short-circuited, the current becomes infinite; that's why real voltage transformers are protected by fuses or similar devices. Now for the current transformer, it is driven by a source that provides a constant current no matter what the load. The transformer takes this current and converts it to some other current depending on the turns ratio; Iout = Iin * Tp / Ts (note the inversion of the turns ratio). For example, if the primary has 1 turn (a common number for real transformers) and the secondary has 5 turns and the primary is supplied with 5 amps, the secondary will provide 1 amp. As the secondary load changes, this current remains the same but the voltage changes. If the secondary is short-circuited, the current still stays the same. If the secondary is open-circuited, the voltage becomes infinite; that's why real portable current transformers have a shorting switch on the secondary that the operator must close before disconnecting the load. Also, note the difference in the number of turns, voltage transformers have a lot of turns and current transformers have few turns. For a loop antenna with an external resonating capacitor, a voltage transformer would be connected in parallel with the loop and capacitor; all three in a parallel circuit. A current transformer would be connected in series with the antenna and capacitor so that the three form a series circuit. If the loop itself is used as the primary of the transformer and another winding is used as the secondary, the distinction between the two types is blurred. Also, a real antenna is neither a voltage source nor a current source but something in between. -- Jim Mueller To get my real email address, replace wrongname with dadoheadman. Then replace nospam with fastmail. Lastly, replace com with us. |
Antenna Simulation in LTspice
On Sun, 10 Mar 2013 17:09:45 -0400, rickman wrote:
On 3/10/2013 1:32 AM, Jeff Liebermann wrote: Holy crap! That's a lot of trouble to see a signal. By "see" I assume you mean on the scope. How large was the signal? Turning off the house was easier than finding the multiple sources of noise at 60KHz. What drove me nuts for about an hour was that much of the noise was coming from my bench oscilloscope. Argh. This is typical. WWVH through an active preamp showing the effect of power line noise (probably from attached switching power supplies). http://www.prc68.com/I/Images/AMRAD110.GIF and after adding some better line filtering: http://www.prc68.com/I/Images/AMRAD_BT.GIF Main page: http://www.prc68.com/I/LF-Ant.shtml I didn't log the setup or take pictures. So, let's do the math and guesswork. http://vk1od.net/calc/FS2RPCalc.htm I plugged in some guesses and recollections as to what the antenna (Q=30) and amp (+20dB gain) were doing and got: http://802.11junk.com/jeffl/crud/WWVH-rx-signal-estimate.jpg -15.8dBm or about 36mv into 50 ohms. I amplified this about 20dB with two or three U310 JFET's (I forgot what I did) to about 3V rms on the scope. I didn't bother with the 50 ohm to scope input Z conversion. Most of what I saw was noise, noise, and more noise. However, if I was patient, I could see the data fade in an out. As I vaguely recall, it was less than 1 division or about 0.1v change. The place where I am working currently is not very close to much and there isn't much in the house. I'm told the fridge is the biggest source of noise. We'll see how the CFL lamps do. Sigh. Most of what I found at 60KHz was coming from lightning storms over Florida. The local sources were all switching power supplies, including those in my test equipment. I didn't have an CFL or LED room lights at the time. I've recently found them to be a rather nasty noise source. Also, the switching power supply wall warts were rather awful. My standard test is to fire up my antique IC-735 HF xceiver, attach a long length of RG-58c/u to the antenna with a resonant loop at the end, tune it to 100KHz (as low as it will go), and sniff around the house. What you'll see on a spectrum analyzer. http://www.prc68.com/I/Spec_0002.shtml If you're thinking of removing all that junk with a 5Hz wide digital filter in software, please note that you'll need to have the input A/D handle the total power of almost all that junk. Also, the amplifier that you're trying to avoid between the antenna and A/D will also need to be rather linear, and therefore rather high power, in order to avoid producing more spurious junk via intermodulation products. Funny, last night my two RCC's both updated like they should. One is an analog clock and runs at 8x speed to get the hour ahead. In the fall it does this to go 11 hours ahead. Quite a sight! They both did the job, but my PC didn't update until it had been on for awhile, without being connected to the I'net. So that's how they change daylight savings time. If I had known, I would have stayed and watched. Thanks for the tip. On the east coast, besides a weak signal, you also have the potential for 60KHz interference from the UK: http://en.wikipedia.org/wiki/MSF_time_signal Loop antennas have a null that can be steered toward the source of interference. I expect that will solve that problem... The depth of the notch seems to be less as the antenna shrinks in size. I'm not sure about this as I haven't attempted to recently model a 60KHz magnetic loop with 4NEC2, but that's what my tinkering shows. If there were a deep notch, most of the home "atomic clock" receivers would be orientation sensitive and I would expect warnings in the docs. The BPSK signal is much better at rejecting interference and digging the signal out of the noise. I don't know exactly how much, but I'm sure it's in a NIST publication somewhere. That's for an ideal receiver. I have my limitations and I have no idea how that will impact the reception. Well, you have to start somewhere, and an ideal receiver is a good place to start. The advantage is that reality only makes everything worse, never better. You should be able to build the BPSK demodulator, and then use a PC to decode the data. I've seen several such programs that do not require I/Q outputs. Here's one based on FreeBSD intended to sync the system clock to WWV/WWVH: http://docs.freebsd.org/doc/4.0-RELEASE/usr/share/doc/ntp/driver36.htm I'm sure there are others. In my case I am not worried that the SNR isn't better, I just need a strong enough signal to drive the LVDS input. Gain at 60KHz is very cheap. Watch out for overload issues. If you design it to work at full scale with whatever you get at 50uV/m, and the signal climbs to 100uV/m, your input A/D isn't going to be very happy. AGC will help, but I don't think it will be needed if you calculate your signal levels so that the A/D input amp isn't clipping. Out of service for a day. It seems that about 30 years of chemistry experiments has finally destroyed much of the kitchen sink plumbing. I hate plumbing. -- Jeff Liebermann 150 Felker St #D http://www.LearnByDestroying.com Santa Cruz CA 95060 http://802.11junk.com Skype: JeffLiebermann AE6KS 831-336-2558 |
Antenna Simulation in LTspice
rickman wrote:
Funny, last night my two RCC's both updated like they should. One is an analog clock and runs at 8x speed to get the hour ahead. In the fall it does this to go 11 hours ahead. Quite a sight! They both did the job, but my PC didn't update until it had been on for awhile, without being connected to the I'net. In Europe we have DCF-77 which is at 77.5 kHz. The trouble receiving it is similar to WWVB. I have several clocks in the house but some of them have only very weak sync. Also, to save battery they only sync once every 12 hours or so. At DST change, they may display the wrong time for a couple of days, especially the one in the kitchen. I need to relocate it to a place where I know there is better signal. The problem is (harmonics of) switching power supplies here as well. Once I had a big open-frame SMPS that I used to power my radio equipment and that switched around 25 kHz. Under the right circumstances, the 3rd harmonic wiped away all DCF-77 receiving within 5 meters or so. Old CRT computer monitors also were problematic. I presume you have some specific needs, low power being among them, to stay focussed on WWVB for your clock sync. Most computer users would use GPS now, or simply sync via the internet. GPS has a different receiving conditions problem, but at least it isn't so much affected by prominently present local interference. Of course a GPS receiver requires a lot more power than a WWVB receiver, although this has come down over the years. Using some sort of on/off switching (syncing with the received time and then having it run free for some time) may help a bit, the battery powered radio clocks do that as well. |
Antenna Simulation in LTspice
On 3/10/2013 9:25 PM, Jim Mueller wrote:
On Fri, 08 Mar 2013 16:51:57 -0500, rickman wrote: On 3/8/2013 4:30 PM, Tim Williams wrote: wrote in message ... Is this a current transformer or a voltage transformer? .--------. .--------. | | | | | C||C VAC C||C Load | C||C | | | | `--------' `--------' Voltage. How about this? .--------. .--------. | | | | | C||C IAC C||C Load | C||C | | | | `--------' `--------' Tim I have to say I don't follow the distinction. It is a transformer, no? The second one is a current transformer. They both consist of coils around a magnetic core driving some kind of load. The difference is the source of power and that causes them to behave very differently as well as being constructed differently. I can't say I understand the distinction. Let's assume ideal components (a good place to start when learning a new concept). The voltage transformer is driven by a source that provides a constant voltage, no matter what the load. The transformer takes this voltage and converts it to some other voltage depending on the turns ratio; Vout = Vin * Ts / Tp. For example, if the primary has 100 turns and the secondary has 20 turns and the primary is supplied with 50 volts, the secondary will provide 10 volts. As the secondary load changes, this voltage remains the same but the current changes. If the secondary is open-circuited, the voltage still stays the same. If the secondary is short-circuited, the current becomes infinite; that's why real voltage transformers are protected by fuses or similar devices. This is ok so far. Now for the current transformer, it is driven by a source that provides a constant current no matter what the load. The transformer takes this current and converts it to some other current depending on the turns ratio; Iout = Iin * Tp / Ts (note the inversion of the turns ratio). For example, if the primary has 1 turn (a common number for real transformers) and the secondary has 5 turns and the primary is supplied with 5 amps, the secondary will provide 1 amp. As the secondary load changes, this current remains the same but the voltage changes. If the secondary is short-circuited, the current still stays the same. If the secondary is open-circuited, the voltage becomes infinite; that's why real portable current transformers have a shorting switch on the secondary that the operator must close before disconnecting the load. I don't follow how any of this has to do with a difference in the transformers. Bth transformers obey both equations you have presented. Both transformers change the voltage as well as the current, no? Also, note the difference in the number of turns, voltage transformers have a lot of turns and current transformers have few turns. I don't see how this follows from what you have written. What is there about these two transformers that define the number of turns? The current transformers I am interested in using use 100 or 300 turns. Is that a lot or just a few? For a loop antenna with an external resonating capacitor, a voltage transformer would be connected in parallel with the loop and capacitor; all three in a parallel circuit. A current transformer would be connected in series with the antenna and capacitor so that the three form a series circuit. If the loop itself is used as the primary of the transformer and another winding is used as the secondary, the distinction between the two types is blurred. Also, a real antenna is neither a voltage source nor a current source but something in between. What I have gotten from this is that Tim's original usage of the terms implies how the transformer is connected to the antenna. As you say, a voltage transformer will be connected across the coil in parallel with the capacitor and a current transformer will be connected in series with the antenna and capacitor. I was planning to use the antenna wire itself in the middle of the antenna loop as the primary of the transformer. So I guess that will be a current transformer. I may try a simulation to see just what happens with a parallel connection. -- Rick |
Antenna Simulation in LTspice
On 3/10/2013 11:42 PM, Jeff Liebermann wrote:
On Sun, 10 Mar 2013 17:09:45 -0400, wrote: On 3/10/2013 1:32 AM, Jeff Liebermann wrote: Holy crap! That's a lot of trouble to see a signal. By "see" I assume you mean on the scope. How large was the signal? Turning off the house was easier than finding the multiple sources of noise at 60KHz. What drove me nuts for about an hour was that much of the noise was coming from my bench oscilloscope. Argh. Yes, like I said, not much in this house and there is not much near it. I have a laptop and my roommate (when he is here) uses one along with an iPhone. I suppose they might generate some noise, but he turns off his laptop at night I'm sure. Otherwise, there just isn't much in the house that isn't 10 or 15 years old. I have a car radio on a linear regulator and an electric shaver that sits charging (part of the time). Otherwise it should be pretty quiet electrically here. This is typical. WWVH through an active preamp showing the effect of power line noise (probably from attached switching power supplies). http://www.prc68.com/I/Images/AMRAD110.GIF and after adding some better line filtering: http://www.prc68.com/I/Images/AMRAD_BT.GIF Main page: http://www.prc68.com/I/LF-Ant.shtml I didn't log the setup or take pictures. So, let's do the math and guesswork. http://vk1od.net/calc/FS2RPCalc.htm I plugged in some guesses and recollections as to what the antenna (Q=30) and amp (+20dB gain) were doing and got: http://802.11junk.com/jeffl/crud/WWVH-rx-signal-estimate.jpg -15.8dBm or about 36mv into 50 ohms. I amplified this about 20dB with two or three U310 JFET's (I forgot what I did) to about 3V rms on the scope. I didn't bother with the 50 ohm to scope input Z conversion. Most of what I saw was noise, noise, and more noise. However, if I was patient, I could see the data fade in an out. As I vaguely recall, it was less than 1 division or about 0.1v change. I'm not sure why you used 1900 meters for the distance. I also don't get why you used 5 ohms for the receiver input impedance. The place where I am working currently is not very close to much and there isn't much in the house. I'm told the fridge is the biggest source of noise. We'll see how the CFL lamps do. Sigh. Most of what I found at 60KHz was coming from lightning storms over Florida. The local sources were all switching power supplies, including those in my test equipment. I didn't have an CFL or LED room lights at the time. I've recently found them to be a rather nasty noise source. Also, the switching power supply wall warts were rather awful. My standard test is to fire up my antique IC-735 HF xceiver, attach a long length of RG-58c/u to the antenna with a resonant loop at the end, tune it to 100KHz (as low as it will go), and sniff around the house. There's just not much of that in this house. What you'll see on a spectrum analyzer. http://www.prc68.com/I/Spec_0002.shtml If you're thinking of removing all that junk with a 5Hz wide digital filter in software, please note that you'll need to have the input A/D handle the total power of almost all that junk. Also, the amplifier that you're trying to avoid between the antenna and A/D will also need to be rather linear, and therefore rather high power, in order to avoid producing more spurious junk via intermodulation products. What A/D? Oh, you mean the LVDS input. How do you saturate a 1 bit ADC? Funny, last night my two RCC's both updated like they should. One is an analog clock and runs at 8x speed to get the hour ahead. In the fall it does this to go 11 hours ahead. Quite a sight! They both did the job, but my PC didn't update until it had been on for awhile, without being connected to the I'net. So that's how they change daylight savings time. If I had known, I would have stayed and watched. Thanks for the tip. On the east coast, besides a weak signal, you also have the potential for 60KHz interference from the UK: http://en.wikipedia.org/wiki/MSF_time_signal Loop antennas have a null that can be steered toward the source of interference. I expect that will solve that problem... The depth of the notch seems to be less as the antenna shrinks in size. I'm not sure about this as I haven't attempted to recently model a 60KHz magnetic loop with 4NEC2, but that's what my tinkering shows. If there were a deep notch, most of the home "atomic clock" receivers would be orientation sensitive and I would expect warnings in the docs. I wouldn't say the *depth* of the null depends on the size of the loop. I think it is a null with a Q, much like a resonance peak, but a null of course. The smaller the loop, the sharper the null like a high Q resonance, so the orientation becomes very critical. In theory at least, the null is perfect, 0 signal. The BPSK signal is much better at rejecting interference and digging the signal out of the noise. I don't know exactly how much, but I'm sure it's in a NIST publication somewhere. That's for an ideal receiver. I have my limitations and I have no idea how that will impact the reception. Well, you have to start somewhere, and an ideal receiver is a good place to start. The advantage is that reality only makes everything worse, never better. You should be able to build the BPSK demodulator, and then use a PC to decode the data. I've seen several such programs that do not require I/Q outputs. Here's one based on FreeBSD intended to sync the system clock to WWV/WWVH: http://docs.freebsd.org/doc/4.0-RELEASE/usr/share/doc/ntp/driver36.htm I'm sure there are others. PC?!!! We don't need no stinking PCs! The demodulator is simple. The signal is beat with a quadrature reference which will bring it down to 0 Hz. This gives two values, a sin and a cos signal. Take the ratio and do an arcTan. This is a simple table lookup made simpler by some convenient math relations. For example, the table only needs to cover 0° to 45° since the ratio can be swapped for 45° to 90° and the other three quadrants distinguished by the sign bits. Some folks would like you to think this has to be done like a high fidelity receiver, but it only has to pull the signal out of the noise. In my case I am not worried that the SNR isn't better, I just need a strong enough signal to drive the LVDS input. Gain at 60KHz is very cheap. Watch out for overload issues. If you design it to work at full scale with whatever you get at 50uV/m, and the signal climbs to 100uV/m, your input A/D isn't going to be very happy. AGC will help, but I don't think it will be needed if you calculate your signal levels so that the A/D input amp isn't clipping. If this is *signal* strength then it won't matter. If this is noise you are talking about, I'm not sure it will be a problem, the signal will still be able to be dug out with enough processing gain. Out of service for a day. It seems that about 30 years of chemistry experiments has finally destroyed much of the kitchen sink plumbing. I hate plumbing. Not much fun, but then what it if you have to work on your knees and get dirty? I don't enjoy working on my car anymore either. -- Rick |
Antenna Simulation in LTspice
On 3/11/2013 5:54 AM, Rob wrote:
wrote: Funny, last night my two RCC's both updated like they should. One is an analog clock and runs at 8x speed to get the hour ahead. In the fall it does this to go 11 hours ahead. Quite a sight! They both did the job, but my PC didn't update until it had been on for awhile, without being connected to the I'net. In Europe we have DCF-77 which is at 77.5 kHz. The trouble receiving it is similar to WWVB. I have several clocks in the house but some of them have only very weak sync. Also, to save battery they only sync once every 12 hours or so. At DST change, they may display the wrong time for a couple of days, especially the one in the kitchen. I need to relocate it to a place where I know there is better signal. Yes, noise can be a problem I understand. I am hoping to get the bandwidth down much more than most receivers so the noise won't be so big a factor. With a signal bandwidth of a handful of Hz, it should be possible. The problem is (harmonics of) switching power supplies here as well. Once I had a big open-frame SMPS that I used to power my radio equipment and that switched around 25 kHz. Under the right circumstances, the 3rd harmonic wiped away all DCF-77 receiving within 5 meters or so. Old CRT computer monitors also were problematic. I presume you have some specific needs, low power being among them, to stay focussed on WWVB for your clock sync. Most computer users would use GPS now, or simply sync via the internet. GPS has a different receiving conditions problem, but at least it isn't so much affected by prominently present local interference. Yes, this is actually a demo to illustrate how low power an FPGA can be. An FPGA will run both the clock and the receiver and use power from the environment rather than batteries. Of course a GPS receiver requires a lot more power than a WWVB receiver, although this has come down over the years. Using some sort of on/off switching (syncing with the received time and then having it run free for some time) may help a bit, the battery powered radio clocks do that as well. Yes, the receiver itself only has to run part of the time, 10% perhaps. The clock has to run 100% obviously. Interesting enough, the FPGA has a base power consumption (0 Hz) of nearly 50% of the power budget and I am confident it will still make the goal. -- Rick |
Antenna Simulation in LTspice
On Mon, 11 Mar 2013 18:10:23 -0400, rickman wrote:
So, let's do the math and guesswork. http://vk1od.net/calc/FS2RPCalc.htm I plugged in some guesses and recollections as to what the antenna (Q=30) and amp (+20dB gain) were doing and got: http://802.11junk.com/jeffl/crud/WWVH-rx-signal-estimate.jpg -15.8dBm or about 36mv into 50 ohms. I amplified this about 20dB with two or three U310 JFET's (I forgot what I did) to about 3V rms on the scope. I didn't bother with the 50 ohm to scope input Z conversion. Most of what I saw was noise, noise, and more noise. However, if I was patient, I could see the data fade in an out. As I vaguely recall, it was less than 1 division or about 0.1v change. I'm not sure why you used 1900 meters for the distance. I also don't get why you used 5 ohms for the receiver input impedance. (Quick reply... still working on my expanding plumbing problem). The 1900 meters is because I screwed up. It should be about 1900Km from San Francisco to Denver. However, any distance greater than zero will suffice for this calculation. The controlling numbers are the 100uV/m field strength, the -3dB antenna gain, and the receiver bandwidth (5Hz). All of the other numbers can change without having any effect on the recovered power. The 5 ohms rx input Z was because the original antenna that I used, was a base loaded 100ft "whip" antenna with a rather impedance. I couldn't decide if the field strength to receive power form wanted the antenna impedance before the 50 ohm matching network, or if it treated the matching as part of the antenna. I flipped a coin and chose 5 ohms. I guess for a loop, 100-200 ohms would be more appropriate. Again, the value makes no difference in the calculations. (Back to plumbing and fixing the 48" farm jack that my neighbor borrowed and returned looking like a pretzel). -- Jeff Liebermann 150 Felker St #D http://www.LearnByDestroying.com Santa Cruz CA 95060 http://802.11junk.com Skype: JeffLiebermann AE6KS 831-336-2558 |
Antenna Simulation in LTspice
Jeff Liebermann wrote: (Back to plumbing and fixing the 48" farm jack that my neighbor borrowed and returned looking like a pretzel). It sounds like your neighbor doesn't know jacks. ;-) |
Antenna Simulation in LTspice
On Mon, 11 Mar 2013, rickman wrote:
Yes, this is actually a demo to illustrate how low power an FPGA can be. An FPGA will run both the clock and the receiver and use power from the environment rather than batteries. But this is all relative. My first "atomic clock" ran on the same set of AA batteries for five or six year, and the second set is now four years old. The Centrios "atomic" wall clock (digital) uses one AA battery, I'm not sure how long that's been in. My watch is a Casio Waveceptor, 3 or 4 years old, which has a solar cell to refresh the battery and it's never been less than fully charged. Fice or six years seems almost as good as 'shelf life" and while I had a small LCD clock (not "atomic") that seemed to run a long time on an AA cell, I'm not sure it was all that lower current than the 'atomic clock". I'm more impressed by the clock running off 1.5v than that the batteries last reasonably long. These things are amazing, considering the effort people used to put into making WWVB receivers, admittedly the "atomic clock" craze has very much benefitted from the power increase at the station. I do orient them, but here in Montreal it's the rare night that they don't sync up, and I don't think since I've had more than one that they all miss the sync. Michael |
Antenna Simulation in LTspice
On Mon, 11 Mar 2013 07:41:41 +0000, Jef wrote:
The second one is a current transformer. They both consist of coils around a magnetic core driving some kind of load. The difference is the source of power and that causes them to behave very differently as well as being constructed differently. What you are describing is a difference in the source NOT a difference in the transformer! The transformer behaves exactly the same in both cases, the inversion is merely down to ohms law. A transformer is a transformer. The only time it gets tricky is when the core and saturation etc etc comes into play. Jeff Very true. A transformer is a transformer and all obey the same equations. But that doesn't prevent real transformers from being called voltage or current transformers. And it doesn't prevent them from being constructed differently or their outputs behaving differently. Part of the problem is that real-world AC current sources are rather rare. So current transformers are seldom connected as shown in the example that was given. They are usually connected in series between a voltage source and some other load, frequently for the purpose of measuring the current. This is what gives rise to the different constructions. Voltage transformers have to withstand the source voltage across their primaries so they have to have enough turns to prevent the core from saturating. The actual number of turns depends on the voltage, frequency, and core size and material. Current transformers are generally constructed to have minimum primary voltage drop so they have few turns on the primary, frequently only one. Considerations of frequency, core size and material still apply, of course. The differences in the name and construction are determined by the intended use. Many current transformers consist of a secondary winding on a toroidal core. The primary is supplied by the user by passing a wire through the hole in the core; thus, one turn. For a loop antenna with a secondary, there is voltage across the antenna winding and current through it. Which is the secondary responding to? The current, since this is what produces the magnetic field. But, since this current is flowing through some number of turns and developing a voltage, it could also be considered a voltage transformer, which is why I said the distinction is blurred. -- Jim Mueller To get my real email address, replace wrongname with dadoheadman. Then replace nospam with fastmail. Lastly, replace com with us. |
Antenna Simulation in LTspice
"Jef" wrote in message
... For a loop antenna with a secondary, there is voltage across the antenna winding and current through it. Which is the secondary responding to? Since you can't have a current thorough the antenna without a voltage the answer is both, It is a chicken and an egg situation tied together by Ohm's Law. That said, low impedances look like constant voltage sources or shorts, and carry a lot of current, while high impedances look like constant current sources or opens, and develop a lot of voltage. It's plenty of an approximation, but when you call an SWR 1.2 a good match, it doesn't take much in either direction to produce that. This is relative to system impedance (50 ohms, say), or if not in a system with characteristic impedances, then some factor of Zo (377 ohms). At "lumped constant" frequencies, Zo and SWR and stuff aren't all that relevant anymore, but the concepts always apply. Tim -- Deep Friar: a very philosophical monk. Website: http://seventransistorlabs.com |
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